Signal detection in frequency division modulated touch systems

ABSTRACT

A frequency division modulated touch detector having row and column conductors arranged such that the path of the row conductors cross the paths of the column conductors, and signal emitters associated with each row, the emitters being adapted to transmit a signal having a specific frequency and initial phase on each row conductor, and a receiver associated with each column to receive signals present on the column conductor. A signal processor is adapted to determine an in-phase and a quadrature component for each of the transmitted signal found in the received signals, and to project a vector representing the transmitted frequencies at their initial phase onto the respective in-phase and quadrature component to determine a measurement for each transmitted signal on each column, and create a heat map reflecting those measurements, the heat map thus containing data reflective of touch.

This application is a continuation of U.S. patent application Ser. No.15/199,395 filed Jun. 30, 2016, which is a non-provisional of and claimsthe benefit of U.S. Provisional Patent Application No. 62/336,150 filedMay 13, 2016, the entire disclosures of each of which are incorporatedherein by reference.

FIELD

The disclosed system and method relate in general to the field of userinput, and in particular to improved signal detection in frequencydivision modulated touch systems.

BRIEF DESCRIPTION OF THE DRAWINGS

The foregoing and other objects, features, and advantages of thedisclosure will be apparent from the following more particulardescription of embodiments as illustrated in the accompanying drawings,in which reference characters refer to the same parts throughout thevarious views. The drawings are not necessarily to scale, emphasisinstead being placed upon illustrating principles of the disclosedembodiments.

FIG. 1 provides a high level block diagram illustrating an embodiment ofa low-latency touch sensor device.

FIG. 2 shows a block diagram illustrating a field flattening procedure.

FIG. 3 shows the relationship between the in-phase and quadraturerepresentation and the amplitude and phase representation.

FIG. 4 shows the signal d corrupted by noise and interference {rightarrow over (n)}.

FIG. 5 shows total noise is divided into components that are paralleland perpendicular to the uncorrupted signal.

DETAILED DESCRIPTION

This application relates to user interfaces such as the fast multi-touchsensors and other interfaces disclosed in U.S. patent application Ser.No. 14/046,819 filed Oct. 4, 2013 entitled “Hybrid Systems And MethodsFor Low-Latency User Input Processing And Feedback,” U.S. patentapplication Ser. No. 13/841,436 filed Mar. 15, 2013 entitled“Low-Latency Touch Sensitive Device,” U.S. Patent Application No.61/798,948 filed Mar. 15, 2013 entitled “Fast Multi-Touch Stylus,” U.S.Patent Application No. 61/799,035 filed Mar. 15, 2013 entitled “FastMulti-Touch Sensor With User-Identification Techniques,” U.S. PatentApplication No. 61/798,828 filed Mar. 15, 2013 entitled “FastMulti-Touch Noise Reduction,” U.S. Patent Application No. 61/798,708filed Mar. 15, 2013 entitled “Active Optical Stylus,” U.S. PatentApplication No. 61/710,256 filed Oct. 5, 2012 entitled “Hybrid SystemsAnd Methods For Low-Latency User Input Processing And Feedback,” U.S.Patent Application No. 61/845,892 filed Jul. 12, 2013 entitled “FastMulti-Touch Post Processing,” U.S. Patent Application No. 61/845,879filed Jul. 12, 2013 entitled “Reducing Control Response Latency WithDefined Cross-Control Behavior,” U.S. Patent Application No. 61/879,245filed Sep. 18, 2013 entitled “Systems And Methods For Providing ResponseTo User Input Using Information About State Changes And PredictingFuture User Input,” U.S. Patent Application No. 61/880,887 filed Sep.21, 2013 entitled “Systems And Methods For Providing Response To UserInput Using Information About State Changes And Predicting Future UserInput,” U.S. patent application Ser. No. 14/046,823 filed Oct. 4, 2013entitled “Hybrid Systems And Methods For Low-Latency User InputProcessing And Feedback,” U.S. patent application Ser. No. 14/069,609filed Nov. 1, 2013 entitled “Fast Multi-Touch Post Processing,” and U.S.Patent Application No. 61/887,615 filed Oct. 7, 2013 entitled “Touch AndStylus Latency Testing Apparatus.” The entire disclosures of thoseapplications are incorporated herein by reference.

Throughout this disclosure, the terms “touch”, “touches,” “contact,”“contacts” or other descriptors may be used to describe events orperiods of time in which a user's finger, a stylus, an object or a bodypart is detected by the sensor. In some embodiments, these detectionsoccur only when the user is in physical contact with a sensor, or adevice in which it is embodied. In other embodiments, the sensor may betuned to allow the detection of “touches” or “contacts” that arehovering a distance above the touch surface or otherwise separated fromthe touch sensitive device. Therefore, the use of language within thisdescription that implies reliance upon sensed physical contact shouldnot be taken to mean that the techniques described apply only to thoseembodiments; indeed, nearly all, if not all, of what is described hereinwould apply equally to “touch” and “hover” sensors. More generally, asused herein, the term “touch” refers to an act that can be detected bythe types of sensors disclosed herein, thus, as used herein the term“hover” is but one type of “touch” in the sense that “touch” is intendedherein. Other types of sensors can be utilized in connection with theembodiments disclosed herein, including a camera, a proximity sensor, anoptical sensor, a turn-rate sensor, a gyroscope, a magnetometer, athermal sensor, a pressure sensor, a force sensor, a capacitive touchsensor, a power-management integrated circuit reading, a keyboard, amouse, a motion sensor, and the like.

The presently disclosed systems and methods provide systems and methodsfor designing, manufacturing and using capacitive touch sensors, andparticularly capacitive touch sensors that employ a multiplexing schemebased on orthogonal signaling such as but not limited tofrequency-division multiplexing (FDM), code-division multiplexing (CDM),or a hybrid modulation technique that combines both FDM and CDM methods.References to frequency herein could also refer to other orthogonalsignal bases. As such, this application incorporates by referenceApplicants' prior U.S. patent application Ser. No. 13/841,436, filed onMar. 15, 2013 entitled “Low-Latency Touch Sensitive Device” and U.S.patent application Ser. No. 14/069,609 filed on Nov. 1, 2013 entitled“Fast Multi-Touch Post Processing.” These applications contemplatecapacitive FDM, CDM, or FDM/CDM hybrid touch sensors which may be usedin connection with the presently disclosed sensors. In such sensors,touches are sensed when a signal from a row is coupled (increased) ordecoupled (decreased) to a column and the result received on thatcolumn.

This disclosure will first describe the operation of fast multi-touchsensors to which the present systems and methods for design,manufacturing and use can be applied. Details of the presently disclosedfrequency division modulated touch system and method are then describedfurther below under the heading “Signal Detection.”

As used herein, the phrase “touch event” and the word “touch” when usedas a noun include a near touch and a near touch event, or any othergesture that can be identified using a sensor. In accordance with anembodiment, touch events may be detected, processed and supplied todownstream computational processes with very low latency, e.g., on theorder of ten milliseconds or less, or on the order of less than onemillisecond.

In an embodiment, the disclosed fast multi-touch sensor utilizes aprojected capacitive method that has been enhanced for high update rateand low latency measurements of touch events. The technique can useparallel hardware and higher frequency waveforms to gain the aboveadvantages. Also disclosed are methods to make sensitive and robustmeasurements, which methods may be used on transparent display surfacesand which may permit economical manufacturing of products which employthe technique. In this regard, a “capacitive object” as used hereincould be a finger, other part of the human body, a stylus, or any objectto which the sensor is sensitive. The sensors and methods disclosedherein need not rely on capacitance. With respect to, e.g., the opticalsensor, such embodiments utilize photon tunneling and leaking to sense atouch event, and a “capacitive object” as used herein includes anyobject, such as a stylus or finger, that that is compatible with suchsensing. Similarly, “touch locations” and “touch sensitive device” asused herein do not require actual touching contact between a capacitiveobject and the disclosed sensor.

FIG. 1 illustrates certain principles of a fast multi-touch sensor 100in accordance with an embodiment. At reference no. 200, a differentsignal is transmitted into each of the surface's rows. The signals aredesigned to be “orthogonal”, i.e., separable and distinguishable fromeach other. At reference no. 300, a receiver is attached to each column.The receiver is designed to receive any of the transmitted signals, oran arbitrary combination of them, with or without other signals and/ornoise, and to individually determine a measure, e.g., a quantity foreach of the orthogonal transmitted signals present on that column. Thetouch surface 400 of the sensor comprises a series of rows and columns(not all shown), along which the orthogonal signals can propagate. In anembodiment, the rows and columns are designed so that, when they are notsubject to a touch event, a lower or negligible amount of signal iscoupled between them, whereas, when they are subject to a touch event, ahigher or non-negligible amount of signal is coupled between them. In anembodiment, the opposite could hold—having the lesser amount of signalrepresent a touch event, and the greater amount of signal represent alack of touch. Because the touch sensor ultimately detects touch due toa change in the coupling, it is not of specific importance, except forreasons that may otherwise be apparent to a particular embodiment,whether the touch-related coupling causes an increase in amount of rowsignal present on the column or a decrease in the amount of row signalpresent on the column. As discussed above, the touch, or touch eventdoes not require a physical touching, but rather an event that affectsthe level of coupled signal.

With continued reference to FIG. 1, in an embodiment, generally, thecapacitive result of a touch event in the proximity of both a row andcolumn may cause a non-negligible change in the amount of signal presenton the row to be coupled to the column. More generally, touch eventscause, and thus correspond to, the received signals on the columns.Because the signals on the rows are orthogonal, multiple row signals canbe coupled to a column and distinguished by the receiver. Likewise, thesignals on each row can be coupled to multiple columns. For each columncoupled to a given row (and regardless of whether the coupling causes anincrease or decrease in the row signal to be present on the column), thesignals found on the column contain information that will indicate whichrows are being touched simultaneously with that column. The quantity ofeach signal received is generally related to the amount of couplingbetween the column and the row carrying the corresponding signal, andthus, may indicate a distance of the touching object to the surface, anarea of the surface covered by the touch and/or the pressure of thetouch.

When a row and column are touched simultaneously, some of the signalthat is present on the row is coupled into the corresponding column (thecoupling may cause an increase or decrease of the row signal on thecolumn). (As discussed above, the term touch or touched does not requireactual physical contact, but rather, relative proximity.) Indeed, invarious implementations of a touch device, physical contact with therows and/or columns is unlikely as there may be a protective barrierbetween the rows and/or columns and the finger or other object of touch.Moreover, generally, the rows and columns themselves are not in touchwith each other, but rather, placed in a proximity that allows an amountof signal to be coupled there-between, and that amount changes(positively or negatively) with touch. Generally, the row-columncoupling results not from actual contact between them, nor by actualcontact from the finger or other object of touch, but rather, by thecapacitive effect of bringing the finger (or other object) into closeproximity—which close proximity resulting in capacitive effect isreferred to herein as touch.

The nature of the rows and columns is arbitrary and the particularorientation is irrelevant. Indeed, the terms row and column are notintended to refer to a square grid, but rather to a set of conductorsupon which signal is transmitted (rows) and a set of conductors ontowhich signal may be coupled (columns). (The notion that signals aretransmitted on rows and received on columns itself is arbitrary, andsignals could as easily be transmitted on conductors arbitrarily namedcolumns and received on conductors arbitrarily named rows, or both couldarbitrarily be named something else.) Further, it is not necessary thatthe rows and columns be in a grid. Other shapes are possible as long asa touch event will touch part of a “row” and part of a “column”, andcause some form of coupling. For example, the “rows” could be inconcentric circles and the “columns” could be spokes radiating out fromthe center. And neither the “rows” nor the “columns” need to follow anygeometric or spatial pattern, thus, for example, the keys on keyboardcould be arbitrarily connected to form rows and columns (related orunrelated to their relative positions.) Moreover, it is not necessaryfor there to be only two types signal propagation channels: instead ofrows and columns, in an embodiment, channels “A”, “B” and “C” may beprovided, where signals transmitted on “A” could be received on “B” and“C”, or, in an embodiment, signals transmitted on “A” and “B” could bereceived on “C”. It is also possible that the signal propagationchannels can alternate function, sometimes supporting transmission andsometimes supporting receipt. It is also contemplated that the signalpropagation channels can simultaneously support transmitters andreceivers—provided that the signals transmitted are orthogonal, and thusseparable, from the signals received. Three or more types of antennaconductors may be used rather than just “rows” and “columns.” Manyalternative embodiments are possible and will be apparent to a person ofskill in the art after considering this disclosure.

As noted above, in an embodiment the touch surface 400 comprised of aseries of rows and columns, along which signals can propagate. Asdiscussed above, the rows and columns are designed so that, when theyare not being touched, one amount of signal is coupled between them, andwhen they are being touched, another amount of signal is coupled betweenthem. The change in signal coupled between them may be generallyproportional or inversely proportional (although not necessarilylinearly proportional) to the touch such that touch is less of a yes-noquestion, and more of a gradation, permitting distinction between moretouch (i.e., closer or firmer) and less touch (i.e., farther orsofter)—and even no touch. Moreover, a different signal is transmittedinto each of the rows. In an embodiment, each of these different signalsare orthogonal (i.e., separable and distinguishable) from one another.When a row and column are touched simultaneously, signal that is presenton the row is coupled (positively or negatively), causing more or lessto appear in the corresponding column. The quantity of the signal thatis coupled onto a column may be related to the proximity, pressure orarea of touch.

A receiver 300 is attached to each column. The receiver is designed toreceive the signals present on the columns, including of any of theorthogonal signals, or an arbitrary combination of the orthogonalsignals, and any noise or other signals present. Generally, the receiveris designed to receive a frame of signals present on the columns, and toidentify the columns providing signal. In an embodiment, the receiver(or a signal processor associated with the receiver data) may determinea measure associated with the quantity of each of the orthogonaltransmitted signals present on that column during the time the frame ofsignals was captured. In this manner, in addition to identifying therows in touch with each column, the receiver can provide additional(e.g., qualitative) information concerning the touch. In general, touchevents may correspond (or inversely correspond) to received signals onthe columns. For each column, the different signals received thereonindicate which of the corresponding rows is being touched in proximitywith that column. In an embodiment, the amount of coupling between thecorresponding row and column may indicate, e.g., the area of the surfacecovered by the touch, the pressure of the touch, etc. In an embodiment,a change in coupling over time between the corresponding row and columnindicates a change in touch at the intersection of the two.

Simple Sinusoid Embodiment

In an embodiment, the orthogonal signals being transmitted onto the rowsmay be unmodulated sinusoids, each having a different frequency, thefrequencies being chosen so that they can be distinguished from eachother in the receiver. In an embodiment, frequencies are selected toprovide sufficient spacing between them such that they can be moreeasily distinguished from each other in the receiver. In an embodiment,frequencies are selected such that no simple harmonic relationshipsexist between the selected frequencies. The lack of simple harmonicrelationships may mitigate non-linear artifacts that can cause onesignal to mimic another.

Generally, a “comb” of frequencies, where the spacing between adjacentfrequencies is constant, and the highest frequency is less than twicethe lowest, will meet these criteria if the spacing between frequencies,Δf, is at least the reciprocal of the measurement period τ. For example,if it is desired to measure a combination of signals (from a column, forexample) to determine which row signals are present once per millisecond(τ), then the frequency spacing (Δf) must be greater than one kilohertz(i.e., Δf>1/τ). According to this calculation, in an example case withonly ten rows, one could use the following frequencies:

Row 1: 5.000 MHz Row 2: 5.001 MHz Row 3: 5.002 MHz Row 4: 5.003 MHz Row5: 5.004 MHz Row 6: 5.005 MHz Row 7: 5.006 MHz Row 8: 5.007 MHz Row 9:5.008 MHz Row 10: 5.009 MHz

It will be apparent to one of skill in the art that frequency spacingmay be substantially greater than this minimum to permit robust design.As an example, a 20 cm by 20 cm touch surface with 0.5 cm row/columnspacing would require forty rows and forty columns and necessitatesinusoids at forty different frequencies. While a once per millisecondanalysis rate would require only 1 KHz spacing, an arbitrarily largerspacing is utilized for a more robust implementation. In an embodiment,the arbitrarily larger spacing is subject to the constraint that themaximum frequency should not be more than twice the lowest (i.e.,f_(max)<2(f_(min))). Thus, in an exemplary embodiment, a frequencyspacing of 100 kHz with the lowest frequency set at 5 MHz may be used,yielding a frequency list of 5.0 MHz, 5.1 MHz, 5.2 MHz, etc. up to 8.9MHz.

In an embodiment, each of the sinusoids on the list may be generated bya signal generator and transmitted on a separate row by a signal emitteror transmitter. In an embodiment, the sinusoids may be pre-generated. Toidentify the rows and columns that are being simultaneously touched, areceiver receives any signals present on the columns and a signalprocessor analyzes the signal to determine which, if any, frequencies onthe list appear. In an embodiment, the identification can be supportedwith a frequency analysis technique (e.g., Fourier transform), or byusing a filter bank. In an embodiment, the receiver receives a frame ofcolumn signals, which frame is processed through an FFT, and thus, ameasure is determined for each frequency. In an embodiment, the FFTprovides an in-phase and quadrature measure for each frequency, for eachframe.

In an embodiment, from each column's signal, the receiver/signalprocessor can determine a value (and in an embodiment an in-phase andquadrature value) for each frequency from the list of frequencies foundin the signal on that column. In an embodiment, where the valuecorresponding to a frequency is greater or lower than some threshold, orchanges from a prior value, that information is used to identify a touchevent between the column and the row corresponding to that frequency. Inan embodiment, signal strength information, which may correspond tovarious physical phenomena including the distance of the touch from therow/column intersection, the size of the touch object, the pressure withwhich the object is pressing down, the fraction of row/columnintersection that is being touched, etc. may be used as an aid tolocalize the area of the touch event. In an embodiment, the determinedvalues are not self-determinative of touch, but rather are furtherprocessed along with other values to determine touch events.

Once values for each of the orthogonal frequencies have been determinedfor at least two frequencies (corresponding to rows) or for at least twocolumns, a two-dimensional map can be created, with the value being usedas, or proportional/inversely proportion to, a value of the map at thatrow/column intersection. In an embodiment, the signals' strengths arecalculated for each frequency on each column. Once signal strengths arecalculated a two-dimensional map may be created. In an embodiment, thesignal strength is the value of the map at that row/column intersection.In an embodiment, values are determined for multiple row/columnintersections on a touch surface to produce a map for the touch surfaceor region. In an embodiment, values are determined for every row/columnintersection on a touch surface, or in a region of a touch surface, toproduce a map for the touch surface or region. In an embodiment, due tophysical differences in the touch surface at different frequencies, thesignal values are normalized for a given touch or calibrated. Similarly,in an embodiment, due to physical differences across the touch surfaceor between the intersections, the signal values need to be normalizedfor a given touch or calibrated.

In an embodiment, touch events are identified using a map produced fromthe value information, and thus, take into account the value changes ofneighboring row/column intersections. In an embodiment, thetwo-dimensional map data may be thresholded to better identify,determine or isolate touch events. In an embodiment, the two-dimensionalmap data may be used to infer information about the shape, orientation,etc. of the object touching the surface.

In an embodiment, such analysis and touch processing described hereinmay be performed on a touch sensor's discrete touch controller. Inanother embodiment, such analysis and touch processing may be performedon other computer system components such as but not limited to one ormore ASIC, MCU, FPGA, CPU, GPU, SoC, DSP or dedicated circuit. The term“hardware processor” as used herein means any of the above devices orany other device (now known or hereinafter developed) which performscomputational functions.

Returning to the discussion of the signals being transmitted on therows, a sinusoid is not the only orthogonal signal that can be used inthe configuration described above. Indeed, as discussed above, any setof signals that can be distinguished from each other will work.Nonetheless, sinusoids may have some advantageous properties that maypermit simpler engineering and more cost efficient manufacture ofdevices which use this technique. For example, sinusoids have a verynarrow frequency profile (by definition), and need not extend down tolow frequencies, near DC. Moreover, sinusoids can be relativelyunaffected by 1/f noise, which noise could affect broader signals thatextend to lower frequencies.

In an embodiment, sinusoids may be detected by a filter bank. In anembodiment, sinusoids may be detected by frequency analysis techniques(e.g., Fourier transform/fast Fourier transform). Frequency analysistechniques may be implemented in a relatively efficient manner and maytend to have good dynamic range characteristics, allowing them to detectand distinguish between a large number of simultaneous sinusoids. Inbroad signal processing terms, the receiver's decoding of multiplesinusoids may be thought of as a form of frequency-divisionmultiplexing. In an embodiment, other modulation techniques such astime-division and code-division multiplexing can also be used. Timedivision multiplexing has good dynamic range characteristics, buttypically requires that a finite time be expended transmitting into (oranalyzing received signals from) the touch surface. Code divisionmultiplexing has the same simultaneous nature as frequency-divisionmultiplexing, but may encounter dynamic range problems and may notdistinguish as easily between multiple simultaneous signals.

As disclosed in U.S. patent application Ser. No. 13/841,436, entitled,“Low-Latency Touch Sensitive Device,” a modulated sinusoid may be usedin lieu of, and as an enhancement of, the simple sinusoid embodimentdescribed above. The entire disclosure of the application isincorporated herein by reference.

Touch surfaces using the previously described techniques may have arelatively high cost associated with generating and detecting sinusoidscompared to other methods. Below are discussed methods of generating anddetecting sinusoids that may be more cost-effective and/or be moresuitable for mass production.

Sinusoid Detection

In an embodiment, sinusoids may be detected in a receiver using acomplete radio receiver with a Fourier Transform detection scheme. Suchdetection may require digitizing a high-speed RF waveform and performingdigital signal processing thereupon. Separate digitization and signalprocessing may be implemented for every column of the surface; thispermits the signal processor to discover which of the row signals are intouch with that column. In the above-noted example, having a touchsurface with forty rows and forty columns, would require forty copies ofthis signal chain. Today, digitization and digital signal processing arerelatively expensive operations, in terms of hardware, cost, and power.It would be useful to utilize a more cost-effective method of detectingsinusoids, especially one that could be easily replicated and requiresvery little power.

In an embodiment, sinusoids may be detected using a filter bank. Afilter bank comprises an array of bandpass filters that can take aninput signal and break it up into the frequency components associatedwith each filter. The Discrete Fourier Transform (DFT, of which the FFTis an efficient implementation) is a form of a filter bank withevenly-spaced bandpass filters that may be used for frequency analysis.DFTs may be implemented digitally, but the digitization step may beexpensive. It is possible to implement a filter bank out of individualfilters, such as passive LC (inductor and capacitor) or RC activefilters. Inductors are difficult to implement well on VLSI processes,and discrete inductors are large and expensive, so it may not be costeffective to use inductors in the filter bank.

At lower frequencies (about 10 MHz and below), it is possible to buildbanks of RC active filters on VLSI. Such active filters may performwell, but may also take up a lot of die space and require more powerthan is desirable.

At higher frequencies, it is possible to build filter banks with surfaceacoustic wave (SAW) filter techniques. These allow nearly arbitrary FIRfilter geometries. SAW filter techniques require piezoelectric materialswhich are more expensive than straight CMOS VLSI. Moreover, SAW filtertechniques may not allow enough simultaneous taps to integratesufficiently many filters into a single package, thereby raising themanufacturing cost.

In an embodiment, sinusoids may be detected using an analog filter bankimplemented with switched capacitor techniques on standard CMOS VLSIprocesses that employs an FFT-like “butterfly” topology. The die arearequired for such an implementation is typically a function of thesquare of the number of channels, meaning that a 64-channel filter bankusing the same technology would require only 1/256th of the die area ofthe 1024-channel version. In an embodiment, the complete receive systemfor the low-latency touch sensor is implemented on a plurality of VLSIdies, including an appropriate set of filter banks and the appropriateamplifiers, switches, energy detectors, etc. In an embodiment, thecomplete receive system for the low-latency touch sensor is implementedon a single VLSI die, including an appropriate set of filter banks andthe appropriate amplifiers, switches, energy detectors, etc. In anembodiment, the complete receive system for the low-latency touch sensoris implemented on a single VLSI die containing n instances of ann-channel filter bank, and leaving room for the appropriate amplifiers,switches, energy detectors, etc.

Sinusoid Generation

Generating the transmit signals (e.g., sinusoids) in a low-latency touchsensor is generally less complex than detection, principally becauseeach row requires the generation of a single signal (or a small numberof signals) while the column receivers have to detect and distinguishbetween many signals. In an embodiment, sinusoids can be generated witha series of phase-locked loops (PLLs), each of which multiply a commonreference frequency by a different multiple.

In an embodiment, the low-latency touch sensor design does not requirethat the transmitted sinusoids are of very high quality, but rather, mayaccommodate transmitted sinusoids that have more phase noise, frequencyvariation (over time, temperature, etc.), harmonic distortion and otherimperfections than may usually be allowable or desirable in radiocircuits. In an embodiment, the large number of frequencies may begenerated by digital means and then employ a relatively coarsedigital-to-analog conversion process. As discussed above, in anembodiment, the generated row frequencies should have no simple harmonicrelationships with each other, any non-linearities in the generationprocess should not cause one signal in the set to “alias” or mimicanother.

In an embodiment, a frequency comb may be generated by having a train ofnarrow pulses filtered by a filter bank, each filter in the bankoutputting the signals for transmission on a row. The frequency “comb”is produced by a filter bank that may be identical to a filter bank thatcan be used by the receiver. As an example, in an embodiment, a 10nanosecond pulse repeated at a rate of 100 kHz is passed into the filterbank that is designed to separate a comb of frequency componentsstarting at 5 MHz, and separated by 100 kHz. The pulse train as definedwould have frequency components from 100 kHz through the tens of MHz,and thus, would have a signal for every row in the transmitter. Thus, ifthe pulse train were passed through an identical filter bank to the onedescribed above to detect sinusoids in the received column signals, thenthe filter bank outputs will each contain a single sinusoid that can betransmitted onto a row.

Transparent Display Surface

It may be desirable that the touch surface be integrated with a computerdisplay so that a person can interact with computer-generated graphicsand imagery. While front projection can be used with opaque touchsurfaces and rear projection can be used with translucent ones, modernflat panel displays (LCD, plasma, OLED, etc.) generally require that thetouch surface be transparent. In an embodiment, the present technique'srows and columns, which allow signals to propagate along them, need tobe conductive to those signals. In an embodiment, the presenttechnique's rows and columns, which allow radio frequency signals topropagate along them, need to be electrically conductive.

If the rows and columns are insufficiently conductive, the resistanceper unit length along the row/column will combine with the capacitanceper unit length to form a low-pass filter: any high-frequency signalsapplied at one end will be substantially attenuated as they propagatealong the poor conductor.

Visually transparent conductors are commercially available (e.g.,indium-tin-oxide or ITO), but the tradeoff between transparency andconductivity is problematic at the frequencies that may be desirable forsome embodiments of the low-latency touch sensor described herein: ifthe ITO were thick enough to support certain desirable frequencies overcertain lengths, it may be insufficiently transparent for someapplications. In an embodiment, the rows and/or columns may be formedentirely, or at least partially, from graphene and/or carbon nanotubes,which are both highly conductive and optically transparent.

In an embodiment, the rows and/or columns may be formed from one or morefine wires that block a negligible amount of the display behind them. Inan embodiment, the fine wires are too small to see, or at least toosmall to present a visual impediment when viewing a display behind it.In an embodiment, fine silver wires patterned onto transparent glass orplastic can be used to make up the rows and/or columns. Such fine wiresneed to have sufficient cross section to create a good conductor alongthe row/column, but it is desirable (for rear displays) that such wiresare small enough and diffuse enough to block as little of the underlyingdisplay as appropriate for the application. In an embodiment, the finewire size is selected on the basis of the pixels size and/or pitch ofthe underlying display.

As an example, the new Apple Retina displays comprises about 300 pixelsper inch, which yields a pixel size of about 80 microns on a side. In anembodiment, a 20 micron diameter silver wire 20 centimeters long (thelength of an iPad display), which has a resistance of about 10 ohms, isused as a row and/or column and/or as part of a row and/or column in alow-latency touch sensor as described herein. Such 20 micron diametersilver wire, however, if stretched across a retina display, may block upto 25% of an entire line of pixels. Accordingly, in an embodiment,multiple thinner diameter silver wires may be employed as a column orrow, which can maintain an appropriate resistance, and provideacceptable response with respect to radiofrequency skin depth issues.Such multiple thinner diameter silver wires can be laid in a patternthat is not straight, but rather, somewhat irregular. A random orirregular pattern of thinner wires is likely to be less visuallyintrusive. In an embodiment, a mesh of thin wires is used; the use of amesh will improve robustness, including against manufacturing flaws inpatterning. In an embodiment, single thinner diameter wires may beemployed as a column or row, provided that the thinner wire issufficiently conductive to maintain an appropriate level resistance, andacceptable response with respect to radiofrequency skin depth issues.

As used below, for convenience of description, the terms transmittingconductor and receiving conductor will be used. The transmittingconductor may be a row or column carrying a signal e.g., from a signalgenerator. In this respect, “conductor” as used herein includes not onlyelectrical conductors but other paths on which signals flow. A receivingconductor may be a row or column carrying a signal resulting from thecoupling of a touch event when a touch event occurs in the proximity ofthe receiving conductor, and not carrying the signal resulting from thecoupling of a touch event when no touch event occurs in the proximity ofthe receiving conductor. In an embodiment, a receiver/signal processormeasures one or more quantities related to each of the orthogonaltransmitted signal on a receiving conductor which signals change overtime as a result of coupling (positive or negative) of a touch event.The measuring of the one or more quantities allows for identification ofa touch event. In an embodiment, the receiver/signal processor maycomprise a DSP, a filter bank, or a combination thereof. In anembodiment, the receiver/signal processor is a comb filter providingbands corresponding to the orthogonal signals.

Because any touch event in proximity to a row-column intersection maychange both the row-signal present on the column, and the column-signalpresent on the row, in an embodiment, any signal on a column or row thatdoes not have a corresponding row or column counterpart may be mitigatedor rejected. In an embodiment, a row-signal received at a columnreceiver/signal processor is used in locating or identifying a touchevent if a corresponding column-signal is received at a correspondingrow receiver/signal processor. For example, a detected signal from Row Rin Column C is only considered to be caused by a touch event if ColumnC's transmitted signal is also detected in Row R. In an embodiment,Column C and Row R simultaneously transmit signals that are orthogonalto the other row and column signals, and orthogonal to each other. In anembodiment, Column C and Row R do not simultaneously transmit signals,but rather, each transmits its signal in an allotted time slice. In suchan embodiment, signals only require frequency- or code-orthogonalityfrom other signals transmitted in the same time slice.

As illustrated, in an embodiment, a single signal generator may be usedto generate the orthogonal signals for both the rows and the columns,and a single signal processor may be used to process the receivedsignals from both the rows and the columns. In an embodiment, one signalgenerator is dedicated to generating row signals and a separate signalgenerator is dedicated to generating column signals. In an embodiment, aplurality of signal generators is dedicated to generating row signalsand the same, or a separate plurality of signal generators is dedicatedto generating column signals. Likewise, in an embodiment, one signalprocessor is dedicated to processing row signals and a separate signalprocessor is dedicated to processing column signals. In an embodiment, aplurality of signal processors are dedicated to processing row signalsand the same, or a separate plurality of signal processors are dedicatedto processing column signals.

In an embodiment, each row and each column may be associated with asignal, and the signal associated with each row or column is unique andorthogonal with respect to the signal for every other row or column. Insuch an embodiment, it may be possible to “transmit” all row and columnsignals simultaneously. Where design or other constraints require, orwhere it is desirable to use fewer than one signal per row and column,time division multiplexing may be employed.

As disclosed in U.S. patent application Ser. No. 14/603,104, filed Jan.22, 2015, entitled “Dynamic Assignment of Possible Channels in a TouchSensor,” a system and method enables a touch sensor to reduce oreliminate such false or noisy readings and maintain a highsignal-to-noise ratio, even if it is proximate to interferingelectromagnetic noise from other computer system components or unwantedexternal signals. This method can also be used to dynamicallyreconfigure the signal modulation scheme governing select portions orthe entire surface-area of a touch sensor at a given point in time inorder to lower the sensor's total power consumption, while stilloptimizing the sensor's overall performance in terms of parallelism,latency, sample-rate, dynamic range, sensing granularity, etc. Theentire disclosure of the application is incorporated herein byreference.

Fast Multi-Touch Post Processing

After the signal strengths from each row in each column have beencalculated using, for example, the procedures described above,post-processing is performed to convert the resulting 2-D “heat map,”also referred to as a “matrix,” into usable touch events. In anembodiment, such post processing includes at least some of the followingfour procedures: field flattening, touch point detection, interpolationand touch point matching between frames. The field flattening proceduresubtracts an offset level to remove crosstalk between rows and columns,and compensates for differences in amplitude between particularrow/column combinations due to attenuation. The touch point detectionprocedure computes the coarse touch points by finding local maxima inthe flattened signal. The interpolation procedure computes the finetouch points by fitting data associated with the coarse touch points toa paraboloid. The frame matching procedure matches the calculated touchpoints to each other across frames. Below, each of the four proceduresis described in turn. Also disclosed are examples of implementation,possible failure modes, and consequences, for each processing step.Because of the requirement for very low latency, the processing stepsshould be optimized and parallelized.

A field flattening procedure may be used to reduce systematic issuesthat cause artifacts in each column's received signal strength. In anembodiment, these artifacts may be compensated-for as follows. First,because of cross-talk between the rows and columns, the received signalstrength for each row/column combination will experience an offsetlevel. To a good approximation, this offset level will be constant andcan be subtracted (or added) off.

Second, the amplitude of the signal received at a column due to acalibrated touch at a given row and column intersection will depend onthat particular row and column, mostly due to attenuation of the signalsas they propagate along the row and column. The farther they travel, themore attenuation there will be, so columns farther from the transmittersand rows farther from the receivers will have lower signal strengths inthe “heat map” than their counterparts. If the RF attenuation of therows and columns is low, the signal strength differences may benegligible and little or no compensation will be necessary. If theattenuation is high, compensation may be necessary or may improve thesensitivity or quality of touch detection. Generally, the signalstrengths measured at the receivers are expected to be linear with theamount of signal transmitted into the columns. Thus, in an embodiment,compensation will involve multiplying each location in the heat map by acalibration constant for that particular row/column combination. In anembodiment, measurements or estimates may be used to determine a heatmap compensation table, which table can be similarly used to provide thecompensation by multiplication. In an embodiment, a calibrationoperation is used to create a heat map compensation table. The term“heat map” as used herein does not require an actual map of heat, butrather the term can mean any array of at least two dimensions comprisingdata corresponding to locations.

In an exemplary embodiment, the entire field flattening procedure is asfollows. With nothing touching the surface, first the signal strengthfor each row signal at each column receiver is measured. Because thereare no touches, substantially the entire signal received is due tocross-talk. The value measured (e.g., the amount of each row's signalfound on each column) is an offset level that needs to be subtractedfrom that position in the heat map. Then, with the constant offsetssubtracted, a calibrated touch object is placed at row/columnintersections and the signal strength of that row's signal at thatcolumn receiver is measured. In an embodiment, all row/columnintersections are used for calibration. The signal processor may beconfigured to normalize the touch events to the value of one location onthe touch surface. The location likely to have the strongest signals canbe arbitrarily chosen (because it experiences the least attenuation),i.e., the row/column intersection closest to the transmitters andreceivers. If the calibrated touch signal strength at this location isS_(N) and the calibrated touch signal strength for each row and columnis S_(R,C) then, if each location in the heat map is multiplied by(S_(N)/S_(R,C)), all touch values will be normalized. In an embodiment,calibrated touches may cause the normalized signal strength for anyrow/column in the heat map to be equal to one.

The field flattening procedure parallelizes well. Once the offsets andnormalization parameters are measured and stored—which should only needto be done once (or possibly again at a maintenance interval)—thecorrections can be applied as soon as each signal strength is measured.FIG. 2 illustrates an embodiment of a field flattening procedure.

In an embodiment, calibrating each row/column intersection may berequired at regular or selected maintenance intervals. In an embodiment,calibrating each row/column intersection may be required once per unit.In an embodiment, calibrating each row/column intersection may berequired once per design. In an embodiment, and particularly where,e.g., RF attenuation of the rows and columns is low, calibrating eachrow/column intersection may not be required at all. Moreover, in anembodiment where the signal attenuation along the rows and columns isfairly predictable, it may be possible to calibrate an entire surfacefrom only a few intersection measurements.

If a touch surface does experience a lot of attenuation, the fieldflattening procedure will, at least to some degree, normalize themeasurements, but it may have some side effects. For example, the noiseon each measurement will grow as its normalization constant gets larger.It will be apparent to one of skill in the art, that for lower signalstrengths and higher attenuations, this may cause errors and instabilityin the touch point detection and interpolation processes. Accordingly,in an embodiment sufficient signal strength is provided for the signalundergoing the largest attenuation (e.g., the farthest row/columnintersection). In an embodiment, after the heat map is generated and thefield flattened, touch points can be identified.

Use Duplication of Sensing to Increase the Sensor's Signal-to-NoiseRatio

A touch sensor can also utilize a number of techniques to decrease theinfluence of interference and other noise in the touch sensor. Forexample, in an embodiment for a touch sensor that employs FDM, a touchsensor could use multiple frequencies per row so that, even if thesensor cannot predict which frequency bins will be subject tointerference, then it can measure each row (or column) in multiple waysand gauge the least noisy measurement (or combination of measurements),and then use those.

In cases where it is difficult to decide whether a measurement has beenaffected by interference or not, a touch sensor could employ a votingscheme whereby a voting plurality of measurements, or a similarstatistical method, is used to determine which measurements to throwaway, which to keep and the best way to statistically and mathematicallycombine the ones it keeps to maximize the signal-to-noise+interferenceratio and thereby enhance the user experience. For example, in anembodiment, an FDM touch sensor subject to interference could transmitthree different frequencies on each row, (where the frequencies aresufficiently separated so that interference between them isstatistically unlikely) and measure the results. Then using atwo-out-of-three voting system, the sensor can determine which of thefrequencies has been degraded the most by interference and, eitherremove its measurement from consideration in the final measurement, orcombine the remaining two in a statistically plausible manner (givenwhat the sensor “knows” a priori about the interference and noisestatistics) or include all three and combine them in a statisticallyplausible manner, weighting the influence of each frequency measurementby the statistical likelihood of its degradation by noise andinterference.

Some methods that a touch sensor can employ in this manner include butare not limited to:

-   -   1. Using multiple frequencies per row. These frequencies could        be employed simultaneously or in sequence.    -   2. Transmitting from rows to columns, and from columns to rows,        which can also be combined with the use of multiple frequencies        above or with another combination of modulation schemes.    -   3. Using CDMA on top of FDM, or some combination of modulation        schemes. Here it should be noted that CDMA signals, unlike those        commonly employed by FDM techniques, are fundamentally        “unnatural” and therefore are often more immune than FDM        modulation schemes to a variety of naturally-occurring signals        in a computer system's external environment.        Improved Signal Detection

The presently disclosed frequency division modulated touch system isused for connection with a touch surface. A frequency division modulatedtouch system must determine the power (or amplitude) of received signalsin order to determine whether a touch event has occurred. Power andamplitude have a functional or proportional relationship, meaning thatwhen one changes the other changes in a predictable manner. Power isnormally calculated by taking the sum-of-squares of the real andimaginary components of an FFT at the frequency of interest. Thesum-of-squares operations require two scalar multiplications and anaddition to determine the power of the estimated signal, as well as asubsequent square root operation to determine the amplitude.

It has been discovered that where the phase of the signal of interest isknown, its amplitude can be estimated with higher signal-to-noise ratio(SNR) by projecting the complex FFT output at that frequency along theunit vector that has the known phase of the received signal. Thisdiscovery excludes noise that would normally be included in thesum-of-squares calculation, thereby increasing the SNR of amplitudeestimation. In an embodiment, half of the noise included in thesum-of-squares calculation is excluded, resulting in an SNR improvementof 3 dB. Moreover, the novel method of amplitude estimation may beimplemented using two scalar multiplications and an addition todetermine the amplitude of the estimated signal. Thus, using the novelapproach disclosed herein, both the SNR and the computational efficiencymay be improved.

The measurement of a particular signal at some frequency has two degreesof freedom; it is therefore a vector of length two. This can beexpressed in polar coordinates as amplitude and phase (i.e., themagnitude and angle of the polar vector), but it can also be expressedin Cartesian coordinates, and usually is designated “in-phase” and“quadrature”. See FIG. 3, which shows the relationship between thein-phase and quadrature representation and the amplitude and phaserepresentation.

One of skill in the art will understand that the phase of a sinusoid isarbitrary because it depends on a particular reference point and time.Considering a chosen reference point, a cosine signal will be maximum atthat point at time t=0, and will then decrease in amplitude. A sinesignal will be zero at that point at time t=0 and will then increase inamplitude. A cosine signal is completely in-phase, meaning that itsvector lies along the x-axis. A sine signal is completely quadrature,meaning that its vector lies along the y-axis. A signal with some otherphase will have both in-phase and quadrature components, and its vectorwill lie between the axes.

Conversion between polar (r,φ) and cartesian (x,y) coordinates can bedone as follows:x=r cos(φ) r=hypot(x,y)=√{square root over (x ² +y ²)}y=r sin(φ) Φ=a tan 2(y,x)

If the phase of a signal measurement is known, one of skill in the artcan convert it to a different phase by rotating the vector to thedesired phase. In polar coordinates this is trivial, because the phasedifference can be added to the original phase. In Cartesian coordinates,a rotation matrix can be used to rotate the vector by the appropriatephase difference.

$\begin{bmatrix}b_{x} \\b_{y}\end{bmatrix} = {\begin{bmatrix}{\cos\left( {\Delta\;\phi} \right)} & {\sin\left( {\Delta\;\phi} \right)} \\{- {\sin\left( {\Delta\;\phi} \right)}} & {\cos\left( {\Delta\;\phi} \right)}\end{bmatrix}\begin{bmatrix}a_{x} \\a_{y}\end{bmatrix}}$Where Δϕ is the phase difference by which the vector is rotated.

The output of a discrete Fourier transform, such as the FFT, is complex.Complex numbers are used as a mathematical convenience to expresstwo-dimensional vectors. The real component represents the x or in-phasecomponent of the vectors, and the imaginary component represents the yor quadrature component. Complex numbers can be used in exponentialform, through Euler's formula:e ^(iu)=cos(u)+i sin(u)An amplitude term r along with the phase term u can be included, whichis sometimes referred to as a phasor:re ^(iu) =r cos(u)+ir sin(u)The exponential form provides a method to multiply two of the vectors ina useful way.re ^(iu) ·se ^(iv) =rs e ^(i(u+v))This formula represents that the product of two vectors (or phasors)with amplitudes r and s, respectively, and phase angles u and v,respectively, is a vector with amplitude rs and a phase angle of u+v.

Discrete Fourier transforms have a certain relationship between patternsin their input and patterns in their output. Specifically, withA(f)=DFT(a(t)):

If the input is . . . Then the output is . . . Real, i.e., imag(a(t)) =0 Even/symmetric, i.e., A(f) = A(−f) Imaginary, i.e., real(a(t)) = 0Odd/anti-symmetric, i.e., A(f) = −A(−f) Complex Neither even nor oddEven/symmetric, i.e., a(t) = a(−t) Real, i.e., imag(A(f)) = 0Odd/anti-symmetric, i.e., Imaginary, i.e., real(A(f)) = 0 a(t) = −a(−t)Neither even nor odd Complex

In FDM-based touch systems, such as those taught in U.S. patentapplication Ser. No. 15/099,179 filed 14 Apr. 2016 (the entiredisclosure of which is incorporated herein by reference), the timedomain data a(t) applied to the input of an FFT is real, but there is noconstraint on it being even or odd. When the duplicated part is ignored,the outputs in the frequency domain are even and complex. The outputfrequency bin A(f) contains a real component (which is the in-phase or xcomponent) and an imaginary component (which is the quadrature or ycomponent). The total power in that bin can be computed by using thefollowing:power in A(f)=real(A(t))²+imag(A(f))²The phase, referenced to input bin a(t=0) of the FFT, isphase of A(f)=a tan 2(imag(A(f)),real(A(f)))

Because the touch systems described herein directly supply thetransmitted signal, and because the transmitters and receivers arerunning in lock-step from the same clock, the phase of each row signalshould always be constant as seen in each column receiver. Note that atouch on the touch sensor does effect the coupling between row andcolumn and may have an effect on the phase. This effect is addressedlater in a subsequent section herein.

The Effects of Noise

The receivers are associated with the columns, to receive column signalspresent on the columns. Further, the receivers on the touch systemsdescribed herein receive not only the deliberately transmitted rowsignals, but also noise and interference. The noise and interference isadditive and is independent of the row signals and independent in eachchannel. It can be modeled as follows:{right arrow over (c)}={right arrow over (a)}+{right arrow over (n)}

Where {right arrow over (c)}, the corrupted signal, is the sum of d, thedeliberately generated row signal, and {right arrow over (n)}, the noiseand interference. FIG. 4 shows a snapshot of the signal, {right arrowover (a)}, corrupted by noise and interference {right arrow over (n)}.

The noise will corrupt the signal in two different ways. First, thecomponent of n that is parallel to {right arrow over (a)} causesamplitude noise, i.e., changes in the amplitude of the sinusoidalsignal. Second, the component of {right arrow over (n)} that isperpendicular to {right arrow over (a)} causes phase noise, i.e.,changes in the phase of the sinusoidal signal. Note that, on average,half of the noise energy goes into amplitude noise and the other halfgoes into phase noise.

The above is strictly true only in the case of high signal-to-noiseratio. While the parallel component of {right arrow over (n)} alwaystranslates into amplitude noise, the perpendicular adds mostly phasenoise, but can also add some amplitude noise.

This is readily understood from FIG. 5. In FIG. 5, the total noise isseparated into components that are parallel and perpendicular to theuncorrupted signal. The parallel component adds an error term to theamplitude of the original signal, and so causes “amplitude noise.” Theperpendicular component adds an error term to the phase of the originalsignal, and so causes “phase noise.” Note that the perpendicularcomponent does not exactly follow the radius of constant amplitude,along which the original signal would rotate if its phase changed.Meaning that, unless the perpendicular component is small compared tothe amplitude of the original signal, it will contribute amplitude noiseas well.

If the perpendicular component is small compared to the amplitude of theuncorrupted signal, it will follow the radius of constant amplitude,meaning that it will add phase noise but not amplitude noise. However,if it gets large enough to deviate substantially from the radius ofconstant curvature, then it will also contribute amplitude noise alongwith phase noise.

Implications for FDM Touch Systems

Touch systems that use frequency division multiplexing (FDM), such asFMT, are prone to additive noise and interference as described above.The signal processing chain normally computes an FFT that provides acomplex output at each frequency of interest, and the power can becalculated by taking the sum of the squares of the real and imaginarycomponents. To produce the amplitude, one would have to take the squareroot of the power, which is computationally expensive. Calculating thepower or amplitude throws away all of the phase information. Thesum-of-squares calculation is:c ² ={right arrow over (c)}·{right arrow over (c)}=({right arrow over(a)}+{right arrow over (n)})·({right arrow over (a)}+{right arrow over(n)})={right arrow over (a)}·{right arrow over (a)}+2{right arrow over(a)}·{right arrow over (n)}+{right arrow over (n)}=a ²+2{right arrowover (a)}·{right arrow over (n)}+n ² =a ²+2an cos φ+n ²The magnitude of the sum-of-squares calculation is its square root:c=√{square root over (a ²+2an cos φ+n ²)}Where φ is the phase angle of the instantaneous noise vector, relativeto the phase angle of the uncorrupted signal.

The known-phase calculation is:d=â·({right arrow over (a)}+{right arrow over (n)})=a+n cos φWhere â is the unit vector in the direction of {right arrow over (a)}.The power of the known-phase calculation is its square:d ²=(a+n cos φ)² =a ²+2an cos φ+n ² cos²φ

The power calculations can be directly compared by noting thatn ² =n ² cos² φ+n ² sin²φThusc ² =a ²+2an cos φ+n ² =a ²+2an cos φ+n ² cos² φ+n ² sin²φ

Power Calculations Equations Sum-of-Squares c² = a² + 2an cos φ +n²cos²φ + n²sin²φ Known-Phase d² = a² + 2an cos φ + n²cos²φ

The difference between the two power methods is n² sin²φ, which isalways non-negative. Therefore, the power calculated by the known-phasemethod will always be smaller than that calculated by the sum-of-squaresmethod. Because the difference term contains only noise variables n andφ, and not the signal variable a, the difference is only composed ofnoise and its elimination makes the measurement closer to the true valueof a².

The magnitude calculations can be similarly compared, but the result isless obvious because of the square root.

Magnitude Calculations Equations Sum-of-Squares c = {square root over(a² + 2an cos φ + n²)} Known-Phase d = a + n cos φ

In certain regimes, by introducing a signal-to-noise ratio variable γ,which is equal to n/a, the reciprocal of the signal-noise-ratio or1/SNR, each of the equations becomes:

$c^{2} = {{a^{2} + {2{an}\;\cos\;\varphi} + n^{2}} = {{a^{2}\left( {1 + {2\;\frac{n}{a}\cos\;\varphi} + \left( \frac{n}{a} \right)^{2}} \right)} = {a^{2}\left( {1 + {2\;\gamma\;\cos\;\varphi} + r^{2}} \right)}}}$$d^{2} = {{a^{2} + {2{an}\;\cos\;\varphi} + {n^{2}\cos^{2}\phi}} = {{a^{2}\left( {1 + {2\;\frac{n}{a}\cos\;\varphi} + {\left( \frac{n}{a} \right)^{2}\cos^{2}\varphi}} \right)} = {a^{2}\left( {1 + {2\gamma\;\cos\;\varphi} + {\gamma^{2}\cos^{2}\varphi}} \right)}}}$$\mspace{20mu}{c = {\sqrt{a^{2} + {2{an}\;\cos\;\varphi} + n^{2}} = {a\sqrt{{1 + {2\;\gamma\;\cos\;\varphi} + \gamma^{2}}\;}}}}$$\mspace{20mu}{d = {{a + {n\;\cos\;\varphi}} = {{a\left( {1 + {\frac{n}{a}\cos\;\varphi}} \right)} = {a\left( {1 + {\gamma\;\cos\;\varphi}} \right)}}}}$

The various regimes can be compared. The first of these is the timeaverage, which can be determined by setting all of the sine and cosineterms to zero (but not their powers since they have to first be reducedto a non-power form).

Calculation Complete Time Average SoS Power c² = a² (1 + 2γcos φ + γ²)c² = a²(1 + γ²) = a² + n² KP Power d² = a² (1 + 2γcos φ + γ²cos²φ)$d^{\; 2} = {{a^{2}\left( {1 + {\frac{1}{2}\gamma^{2}}} \right)} = {a^{2} + {\frac{1}{2}n^{2}}}}$Sos Mag c = a {square root over (1 + 2γcos φ + γ²)} c = a {square rootover (1 + γ²)} = {square root over (a² + n²)} KP Mag d = a (1 + γcos φ)d = a

The time average is the baseline of the measured signal, ignoringdeviations from it. The power of the known-phase measurement has onlyhalf the noise contribution of the sum-of-squares measurement, and thushas a 3 dB SNR improvement. The time average of the sum-of-squaresmagnitude has a bias term and will always be larger than the timeaverage of the known-phase measurement.

Note that the known-phase calculation ignores the noise component thatis orthogonal to the known signal, i.e., the n sin ϕ component of {rightarrow over (n)}. That component contains only noise and none of theoriginal signal, so there is zero utility in including it.

In an embodiment, further note that the 3 dB reduction in noise is anaverage, having been integrated over a large number of samples. Anestimate of the original signal amplitude calculated with theknown-phase technique may have just as much noise as an estimate thatuses the sum-of-squares technique (if the noise vector happens to beparallel to the original signal vector), or no noise at all (if thenoise vector happens to be perpendicular to the original signal vector).As long as the original signal phase is truly known, the known-phasetechnique will never produce a result with lower SNR than thesum-of-squares technique.

To determine the deviations from the time-averaged baseline,instantaneous calculations must be used and the sine and cosine termscannot be omitted. Instead, the calculations in both the high- andlow-SNR regimes are used to see how each is affected. Below is anexamination of the high-SNR regime, in which n<<a and therefore γ<<1.

Calculation Complete High SNR SoS Power c² = a²(1 + 2γcos φ + γ²) c² ≈a²(1 + 2γcos φ) KP Power d² = a²(1 + 2γcos φ + γ²cos²φ) d² ≈ a²(1 +2γcos φ) SoS Mag c = a{square root over (1 + 2γcos φ + γ²)} c ≈ a{squareroot over (1 + 2γcos φ)} ≈ a (1 + γcos φ) KP Mag d = a (1 + γcos φ) d ≈a(1 + γcos φ)

Because γ<<1, γ²<<γ all γ² terms can be omitted if there are also γterms. After such omissions, however, there is no advantage of using theknown-phase technique over the sum-of-squares technique, and vice versa.In the high-SNR regime, all of these techniques perform equally.

Below is an examination of the low-SNR regime, in which a<<n andtherefore γ>>1 and γ²>>γ.

Calculation Complete High SNR SoS Power c² = a²(1 + 2γcos φ + γ²) c² ≈n² KP Power d² = a² (1 + 2γcos φ + γ²cos²φ)$d^{2} \approx {\frac{1}{2}{n^{2}\left( {1 + {\cos\mspace{14mu} 2\varphi}} \right)}}$SoS Mag c = a{square root over (1 + 2γcosφ + γ²)} c ≈ n KP Mag d = a(1 +γcos φ) d ≈ n cos φ

In the low-SNR regime, the signal a goes away and leaves only noise. Theknown-phase technique never produces more noise than the sum-of-squarestechnique, even on an instantaneous basis, and on average produces onlyhalf of the noise. On an instantaneous basis, the known-phase techniquewill sometimes produce the same amount of noise as the sum-of-squarestechnique, and sometimes will produce none.

In a real-world application, the SNR will be somewhere between the twoextremes.

Exemplary Embodiment

To estimate the original signal power using a sum-of-squares technique,the complex value of the particular FFT output bin and A(f) are used tocalculate:power estimate=P _(SoS)(f)=real(A(f))²+imag(A(f))²The estimated amplitude is the square root of the power estimate.In an embodiment, to estimate the original signal amplitude using theknown-phase technique, the known phase is required. In anotherembodiment, this information can be obtained from design information, orcan be measured directly from the touch system. In the presence of noiseand interference, it would be best to average or otherwise statisticallycombine many values of the particular FFT output. In an embodiment, thismay be done by averaging the real and imaginary components separately,so the known phase would therefore be:average known phase=φ_(k) =a tan 2(mean(imag(A(f))),mean(real(A(f))))Then the known phase is converted to a unit vector with that phase:û _(k)=[cos φ_(k),sin φ_(k)]Taking the dot product of this unit vector with the incoming complexsamples will yield the amplitude estimates at that frequency:amplitude estimate Y _(KP)(f)=real(A(f))cos ϕ_(k)+imag(A(f))sin ϕ_(k)The corresponding power estimate P_(KP)(f)=(Y_(KP)(f))². Note, as wouldbe understood by one of skill in the art, the dot product operationabove is half of the calculation for multiplication by a rotationmatrix. It will also be apparent to one of skill in the art in view ofthis disclosure that other statistical combinations may be usefulinstead of average, such as, e.g., median, or mode, or other measurethat reflects a characteristic of the values.

It should be noted that no trigonometric functions are required in theembedded system itself. These can be computed beforehand, either atdesign time or at calibration time, if they are needed at all. In fact,no trigonometric functions may be needed at all. The unit vectorû_(k)=[cos ϕ_(k), sin ϕ_(k)] is just the average of the real andimaginary values, which has then been normalized. If it is necessary forthe unit vectors to be computed by the embedded device itself (during apower-on calibration interval, perhaps), the most expensive computationneeded is a division operator for the normalization. The divisionoperator must be done once for each row frequency and then is never usedagain.

Non-Constant Signal Phase

Where the transmitted phase is not constant relative to the receiver, atouch on the sensor causes a phase change between the transmitter andreceiver, or the phase was measured incorrectly, then that could resultin the phase of the original signal not being constant or otherwise notmatching the “known” phase used in the calculation.

Small phase differences make little difference in the finalsignal-to-noise ratio, however. The phase error would affect only theoriginal signal, but not the noise (on average) because the anglebetween the noise and the signal is uncorrelated.

In an embodiment, using the known-phase technique, a phase difference ofΔϕ would lower the amplitude of the received signal by cos Δϕ, andtherefore its power by cos²Δϕ. In another embodiment, a phase differenceof 10 degrees would lower the measured SNR by 0.13 dB. In anotherembodiment, a difference of 30 degrees would lower the SNR by 1.25 dBand a difference of 45 degrees would lower the SNR by 3 dB. On average,use of the known-phase technique will have an SNR advantage as long asthe average phase error does not exceed 45 degrees.

In an embodiment, use of the known-phase technique to compute theestimated amplitude of a transmitted signal will provide an average SNRimprovement of 3 dB, while requiring approximately the same amount ofcomputational resources as the existing sum-of-squares technique. Evenfewer resources are required if amplitude results are preferred overpower results because the amplitude is calculated directly with no needto compute a square root.

In an embodiment, the only additional resources required are themeasurement of the known signal phases, which can be calculated atdesign time or measured after a device is built, and two memory storagelocation per used FFT frequency bin. Each storage location must be ableto hold a scalar with value between −1 and +1.

Any of the resulting measurements can be a measurement of power, ameasurement of amplitude, or is proportional to a measurement ofamplitude.

In an embodiment, the touch detector is comprised of one or more rowsand one or more columns of conductive material, at least one signalemitter, at least one receiver, and at least one signal processor.

In some exemplary embodiments, the rows and columns are arranged in amatrix of rows and columns of conductive material. In anotherembodiment, the touch detector contains first and second row conductors,and a column conductor, arranged such that the path of the first andsecond row conductors cross the path of the column conductor. In anotherembodiment, the touch detector contains at least one first row conductorand at least one first column conductor arranged such that the path ofthe first row conductor crosses the path of the first column conductor.

In another embodiment, a first row conductor and a first columnconductor arranged such that the path of the first row conductor crossesthe path of the first column conductor. Additionally, at least oneadditional row conductor is present, and is arranged such that the pathof the at least one additional row conductor crosses the path of thefirst column conductor. Additionally, the at least one additional rowconductor is a plurality of additional row conductors, and each of theplurality of additional row conductors are arranged such that the pathof each of the plurality of additional row conductors crosses the pathof the first column conductor. In another embodiment, there is at leastone additional column conductor arranged such that the path of the atleast one additional column conductor crosses the path of the first rowconductor and the path of the at least one additional row conductor. Inan embodiment, there is at least one additional column conductor, theone additional column conductor being arranged such that the path of theat least one additional column conductor crosses the path of the firstrow conductor. In another embodiment, the at least one additional columnconductor is a plurality of additional column conductors, and each ofthe plurality of additional column conductors are arranged such that thepath of each of the plurality of additional column conductors crossesthe path of the first row conductor. In another embodiment, the at leastone additional row conductor is arranged such that the path of the atleast one additional row conductor crosses the paths of the first columnconductor and the path of the at least one additional column conductor.

The known-phase technique is compatible with, and may be advantageousfor use in connection with, certain touch sensor technology, includingbut not limited to those various methods and apparatus disclosed in theU.S. patent applications identified in the first paragraph of thisDetailed Description.

The present systems and methods are described above with reference toblock diagrams and operational illustrations of methods and devices forfrequency conversion and heterodyning. It is understood that each blockof the block diagrams or operational illustrations, and combinations ofblocks in the block diagrams or operational illustrations, may beimplemented by means of analog or digital hardware and computer programinstructions. These computer program instructions may be provided to aprocessor of a general purpose computer, special purpose computer, ASIC,or other programmable data processing apparatus, such that theinstructions, which execute via the processor of the computer or otherprogrammable data processing apparatus, implements the functions/actsspecified in the block diagrams or operational block or blocks. In somealternate implementations, the functions/acts noted in the blocks mayoccur out of the order noted in the operational illustrations. Forexample, two blocks shown in succession may in fact be executedsubstantially concurrently or the blocks may sometimes be executed inthe reverse order, depending upon the functionality/acts involved.

While the invention has been particularly shown and described withreference to a preferred embodiment thereof, it will be understood bythose skilled in the art that various changes in form and details may bemade therein without departing from the spirit and scope of theinvention.

What is claimed is:
 1. A device, comprising: signal emitter adapted totransmit a unique one of a plurality of signals, each of the pluralityof signals having a respective initial phase and a respective frequency,and each of the signals being frequency orthogonal to each of the otherplurality of signals; receiver to receive signals; and signal processorbeing adapted to: (i) for each received signal, determine an in-phaseand a quadrature component for the respective frequencies of each of theplurality of signals, each pair of in-phase and quadrature componentsdefining a vector reflecting a component of the received signal at therespective frequency; and (ii) determine an estimate for each of therespective frequencies by computing a dot product of a unit vectorrepresenting each signal at its corresponding respective phase and thevector represented by the determined in-phase and quadrature components;and (iii) identify an event based on the estimates.
 2. The deviceaccording to claim 1, wherein the signal processor and the receiver arepart of the same component.
 3. The device according to claim 1, whereinthe signal processor and the receiver are not part of the samecomponent.
 4. The device according to claim 1, wherein the respectiveinitial phase of one of the plurality of signals is different than therespective initial phase of another of the plurality of signals.
 5. Thedevice according to claim 4, wherein the respective initial phase of theone of the plurality of signals is the same as the respective initialphase of yet another of the plurality of signals.
 6. The deviceaccording to claim 1, wherein the respective initial phase of one of theplurality of signals is the same as the respective initial phase ofanother of the plurality of signals.
 7. A device, comprising: conductorarranged such that an event external to the device causes a change incoupling between at least two conductors; first and second signalemitters respectively adapted to transmit a first signal having a firstfrequency at a first phase and a second signal having a second frequencyat a second phase, each of the first and second signals being orthogonalto the other; receiver to receive signals; signal processor beingadapted to: determine an in-phase and a quadrature component of thefirst and second frequencies in the received signals; determine a firstestimate for the first frequency by computing a dot product of a firstvector representing the first frequency at its corresponding first phaseand the vector represented by the determined in-phase and quadraturecomponents of the first frequency; determine a second estimate for thesecond frequency by computing a dot product of a second vectorrepresenting the second frequency at its corresponding second phase andthe vector represented by the determined in-phase and quadraturecomponents of the second frequency; and detect an event based on thefirst and second estimates.
 8. The device according to claim 7, whereinthe signal processor and the receiver are part of the same component. 9.The device according to claim 7, wherein the signal processor and thereceiver are not part of the same component.
 10. The device according toclaim 7, wherein at least one of the first and second estimates is anestimate of power.
 11. The device according to claim 7, wherein at leastone of the first and second estimates is an estimate of amplitude. 12.The device according to claim 7, wherein at least one of the first andsecond estimates is proportional to an estimate of amplitude.
 13. Thedevice according to claim 7, wherein the signal processor comprises morethan one signal processor units, each of the signal processor units-sadapted to perform at least one of the following operations: determinean in-phase and a quadrature component of the first and secondfrequencies in the received signals; determine a first estimate for thefirst frequency by computing a dot product of a first vectorrepresenting the first frequency at its corresponding first phase andthe vector represented by the determined in-phase and quadraturecomponents of the first frequency; determine a second estimate for thesecond frequency by computing a dot product of a second vectorrepresenting the second frequency at its corresponding second phase andthe vector represented by the determined in-phase and quadraturecomponents of the second frequency; and detect an event based on thefirst and second estimates; and all of the one or more signal processorunits, collectively, perform all of the operations.
 14. The deviceaccording to claim 13, wherein at least one of the first and secondestimates is an estimate of power.
 15. The device according to claim 13,wherein at least one of the first and second estimates is an estimate ofamplitude.
 16. The device according to claim 13, wherein at least one ofthe first and second estimates is proportional to an estimate ofamplitude.
 17. A method for detecting an event, comprising: transmittinga first and second signal, the first signal being at a first frequencyand a first phase, and the second signal being at a second frequency anda second phase, wherein the first and second frequencies are orthogonalto each other; receiving signals; determining an in-phase and aquadrature component of the first frequency in the received signal, thein-phase and quadrature components of the first frequency defining afirst vector reflecting a component of the received signal at the firstfrequency; determine a first estimate for the first frequency bycomputing a dot product of a vector representing the first signal at thefirst phase and the first vector; determining an in-phase and aquadrature component of the second frequency in the received signal, thein-phase and quadrature components of the second frequency defining asecond vector reflecting a component of the received signal at thesecond frequency; determine a second estimate for the second frequencyby computing a dot product of a second vector representing the secondsignal at the second phase and the second vector; and creating a matrixreflecting the detected estimates.
 18. The method of claim 17, furthercomprising using the matrix to determine an event.
 19. The method ofclaim 17, further comprising: repeating the transmitting step, thereceiving step, the four determining steps and the creating step, andusing sequential matrices to detect an event.
 20. The method of claim17, wherein the estimate related to the event is an estimate ofamplitude.